PA block

Starting point of the PA design was a 75-μm-wide three-stacked power cell containing also the upper gate capacitors needed for even voltage distribution among the stacked transistors. The stacking enabled increasing the drain bias voltage up to 2.7 V. The main signal driver (DR) and the PA pair forming the output stage (BA1 and BA2) were then assembled by grouping the power cells into 150 μm pseudo-differential branches as in Fig. 2. The width of the PA blocks was chosen to maximize output power with good linearity. Differential topology was chosen for reliability and for the ease of power cell combination and drain supply feeding via center tap of the output balun. The CA can be much smaller than the rest of the amplifier blocks, since control signal levels of −10 dB compared to the main input signal can already cause significant load modulation [Reference Collins, Quaglia, Powell and Cripps27]. CA has half of the width (75 μm per branch) compared to the other PA blocks and has a source degeneration transistor (M0, 100 μm) with back-gate bias option in addition to the normal gate bias $V_{\rm g1}$, allowing better adjustment of its output power. CA output range spans from −27 dB to −4 dB, compared to the driver amplifier output power, and it can be adjusted with 1–2 dB steps. All of the PA blocks follow the same principle; the input matching and transformation from single-ended to pseudo-differential mode is done with a stacked transformer. On the output side, the power matching and transformation from pseudo-differential to single-ended is implemented with a center-tapped transformer, where the center tap functions as the drain bias line.

Figure 2. Schematic of a pseudo-differential PA building block.

Quadrature hybrid

(1)\begin{align} & Y_{\rm Qhyb} =\\ & \begin{bmatrix} \frac{1}{Z_{\rm C1}}\!+\!\frac{Z_{\rm L2}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & -\frac{1}{Z_{\rm C1}}\!-\!\frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & -\frac{Z_{\rm L2}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & \frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} \\ -\frac{1}{Z_{\rm C1}}\!-\!\frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & \frac{1}{Z_{\rm C1}}\!+\!\frac{Z_{\rm L1}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & \frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} &-\frac{Z_{\rm L1}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} \\ -\frac{Z_{\rm L2}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & \frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & \frac{1}{Z_{\rm C2}}\!+\!\frac{Z_{\rm L2}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & -\frac{1}{Z_{\rm C2}}\!-\!\frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} \\ \frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & -\frac{Z_{\rm L1}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & -\frac{1}{Z_{\rm C2}}\!-\!\frac{Z_{\rm M}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2} & \frac{1}{Z_{\rm C2}}\!-\!\frac{Z_{\rm L1}}{Z_{\rm L1}Z_{\rm L2}-Z_{\rm M}^2}\end{bmatrix} \end{align}

The quadrature hybrids in this work are based on a transformer tipped on its side. The coupling is further adjusted with capacitors. Benefits in utilizing transformer-based quadrature hybrids, as opposed to transmission line-based coupler implementations, are compact size and straightforward design approach. Figure 3 shows the quadrature hybrid schematic and frequency response. The general form of the resulting network is shortest to illustrate with a Y-parameter matrix (1). $Z_{\rm L1}$, $Z_{\rm L2}$, and Z _M in Equation (1) are the impedances of the primary side, secondary side, and mutual inductance, respectively. $Z_{\rm C1}$ and $Z_{\rm C2}$ correspondingly are impedances of the tuning capacitors. When the Y-parameter matrix is converted to the S-parameter matrix and we assume that the quadrature hybrid is symmetric ($L_1=L_2$ and $C_1=C_2$), then at a center frequency, the hybrid’s S-parameter matrix can be written as

(2)\begin{equation} S_{{\rm Qhyb},f_{c}} = \begin{bmatrix} 0 & 0 & \frac{1}{\sqrt{2}}{\rm e}^{j\phi_d} & \frac{1}{\sqrt{2}}{\rm e}^{j\phi_c} \\ 0 & 0 & \frac{1}{\sqrt{2}}{\rm e}^{j\phi_c} & \frac{1}{\sqrt{2}}{\rm e}^{j\phi_d} \\ \frac{1}{\sqrt{2}}{\rm e}^{j\phi_d} & \frac{1}{\sqrt{2}}{\rm e}^{j\phi_c} & 0 & 0 \\ \frac{1}{\sqrt{2}}{\rm e}^{j\phi_c} & \frac{1}{\sqrt{2}}{\rm e}^{j\phi_d} & 0 & 0\end{bmatrix}, \end{equation}

that is the hybrid provides good matching and isolation, and direct (S ₁₃) and coupled (S ₁₄) paths are equal in magnitude with phase offset relation $\phi_c=\phi_d+90^\circ$.

Figure 3. (a) Schematic of the transformer-based quadrature hybrid and (b) EM-simulated S-parameter plot.

Load modulation

As established in [Reference Collins, Quaglia, Powell and Cripps27], the OLMBA load modulation is achieved by using the input quadrature hybrid isolation port (i.e. port 2 in Fig. 3, or correspondingly QH Mid. port 2 in Fig. 1) as a second input. The second input RF signal is referred to as the control signal, and in Equations (3) and (4), it is expressed as relative term $\alpha = \frac{V_{\rm in2}}{V_{\rm in1}}$, where $V_{\rm in1}$ and $V_{\rm in2}$ refer to main and control signals, respectively. The control signal is amplified by the balanced PA pair and reflects back to their outputs from a reflective load placed to the output quadrature hybrid isolation port, causing load modulation.

(3)\begin{equation} \Gamma_{\rm BA1} = {{\begin{array}{c}(K_{\rm BA2}-K_{\rm BA1})j\Gamma_{\rm iso}\\+\alpha(K_{\rm BA2}+K_{\rm BA1})\Gamma_{\rm iso}\end{array}}\over{\begin{array}{c}(K_{\rm BA2}-K_{\rm BA1})j\Gamma_{\rm iso}\Gamma_{\rm PA}+2K_{\rm BA1}\\+\alpha((K_{\rm BA2}+K_{\rm BA1})\Gamma_{\rm iso}\Gamma_{\rm PA}+2jK_{\rm BA1})\end{array}}} \end{equation}

(4)\begin{equation} \Gamma_{\rm BA2} = {{\begin{array}{c}(K_{\rm BA2}-K_{\rm BA1})j\Gamma_{\rm iso}\\+\alpha(K_{\rm BA2}+K_{\rm BA1})\Gamma_{\rm iso}\end{array}}\over{\begin{array}{c}(K_{\rm BA2}-K_{\rm BA1})j\Gamma_{\rm iso}\Gamma_{\rm PA}+2K_{\rm BA2}\\+\alpha((K_{\rm BA2}+K_{\rm BA1})\Gamma_{\rm iso}\Gamma_{\rm PA}-2jK_{\rm BA2})\end{array}}} \end{equation}

In this work, the control signal is generated by dividing it from the main signal input with a quadrature hybrid (QH In in Fig. 1) and amplifying it with the CA in a parallel path to the main signal driver amplifier (DR). This is to say that the control signal is fixed to $-90^\circ$ phase shift compared to the main path, except for any bias or frequency-dependent deviations in the phase relationship. The proposed OLMBA is capable of providing the $|\alpha|$ approximately up to −4 dB. The output quadrature hybrid has its isolation port terminated with a large capacitor, which is essentially a short circuit at the operating frequency. This control signal amplitude and phase and output isolation port load choice combination realizes load modulation between maximum and backed off power matching points as illustrated in Fig. 4. Both of the output stage-balanced PAs have their own bias controls, which enables additional degrees of freedom in the load modulation. The general form of the load modulation, expressed as reflection coefficients seen by BA1 and BA2 when antenna port load is assumed to be $Z_0 = 50 \Omega$, is given in Equations (3) and (4), respectively. The equations are divided into two sections: only bias and isolation port load dependent and bias, control signal, and isolation port load dependent. $K_{\rm BA1}$ and $K_{\rm BA2}$ denote the fundamental bias-dependent transconductances and $\Gamma_{\rm iso}$ denotes the output quadrature coupler reflective load. To be exact, the load modulation depends also on the S ₂₂ of the PAs, which is, in this case, assumed to be the same for both PAs and is denoted by $\Gamma_{\rm PA}$.

The load modulation during a power sweep is illustrated with simulations in Fig. 4. The choice of $\Gamma_{\rm iso} \approx -1$ and $\angle\alpha \approx -90^\circ$ causes the BA impedances to move both on a real and imaginary axes, as illustrated in Fig. 4(a). Initially, the control signal pushes the impedances higher toward the back-off efficiency matching point (red square), and they release during the power sweep toward the maximum power matching point (red pentagram). Black and magenta circles in Fig. 4 indicate the endpoints of the power sweep for BA1 (blue curves) and BA2 (green curves), respectively. Figure 4(b), on the other hand, illustrates load modulation during a power sweep, which can be achieved by setting BA1 and BA2 to uneven bias. When BA1 is at a higher bias, the load modulation moves roughly in a similar direction as with load modulation caused by the control signal. When BA2 dominates, the load modulation direction is the opposite. Lastly, Fig. 4(c) shows the ensuing load modulation when all controls are used.

Figure 4. Illustration of load modulation behavior during power sweep depending on (a) DR and CPA bias settings, (b) BA1 and BA2 bias settings, and (c) both utilized simultaneously. The bias settings were swept from 290 mV to 380 mV with 30 mV steps. CA back-gate bias was set to 1.5 V (resulting in $|\alpha|$ in the range of −10 to −7 dB.) Blue and green lines depict impedances in BA1 and BA2 75 µm power cell drain, respectively. Black and magenta circles indicate the endpoints of the power sweep. The red square indicates the optimal impedance for back-off efficiency and the red pentagram is the maximum power efficiency point.

Continuous wave measurements

A micrograph of the fabricated circuit is presented in Fig. 5. Continuous wave (CW) measurements were done using Keysight PNA-X network analyzer (N5247B, 4-port, 67 GHz) and FormFactor Infinity I40 probes. For the power sweeps, the power calibration plane was at the end of the cables and the probe losses (about 0.5 dB) were deducted from the results. The calibration was normalized using an external calibration substrate (Cascade P/N 101-190), and therefore the probe pads are included in the measurements. The chip had programmable DC biases for the lowest gates ($V_{\rm g1}$) of the stacked transistors. Upper gate biases ($V_{\rm g2}$ and $V_{\rm g3}$) were set to fixed values and generated with off-chip voltage supplies, so that none of the transistors would exceed maximum V _ds of 900 mV. The PA bias settings and the measurement equipment were controlled using MATLAB. Block diagram of the CW measurement setup is shown in Fig. 6(a).

Figure 5. Micrograph of the fabricated PA.

Figure 6. Block diagrams of (a) CW measurement setup and (b) modulated signal measurement setup.

Measured and simulated S-parameters are presented in Fig. 7. The measured frequency response (solid lines) has shifted lower compared to the simulated (dashed lines). This is dominantly because the whole structure was already demanding to simulate as a single EM-block, and because of that, the metal fills were not included in the simulations. The measured 3-dB bandwidth ended up spanning from 22.8 GHz to 28.5 GHz, with 26 GHz as the center frequency. S11 and S22 both remain below −10 dB throughout the operating range. S12 stays below −45 dB and is omitted for clarity.

Figure 7. Measured and simulated S-parameters, with same bias settings as with 26 GHz results in Table 1.

For the power sweeps, the biases of DR, CA, BA1 and BA2 were swept with a focus on finding the settings producing the high output 1-dB compression point (P_1dB) with minimal amplitude to phase modulation (AM-PM). The estimated bias range for the ($V_{\rm g1}$) sweeps was from 290 to 380 mV, with cut-off voltage being at 200 mV and saturation voltage at 490 mV. Measurement results at different frequencies with their respective bias settings (for brevity expressed as the total drain current I_dQ) are listed in Table 1. Figure 8 shows an example of a power sweep at 26 GHz, where otherwise the same bias settings are compared with and without control amplifier contribution. When the control amplifier is on, which is the blue curves, we get some penalty in gain, but the compression point improves by one dB and AM-PM improves by five degrees. Additionally, the output stage drain efficiency improves by three percentage points. Figure 8 also contains normalized reference lines for ideal class A and B efficiencies. At 6 dB back-off from $P_{\rm 1\,dB}$, the efficiency is as good as with class B and 2.3 times better than class A.

Table 1. Measurement results using CW and modulated signals

Figure 8. CW power sweep results measured at 26 GHz.

The integrated OLMBA was compared to a stand-alone integrated quadrature balanced amplifier (BA) with 50 Ω loads at the isolation ports, as illustrated in Fig. 9(a). The micrograph of the BA PA is shown in Fig. 9(b). The stand-alone BA test circuit consists of BA1 and BA2 without a driving stage, fed and terminated with GSGSG pads. Power sweep results at 26 GHz are compared in Fig. 10 and in Table 2. Without driver amplifier, the measurement equipment source power was adequate to push the stand-alone BA only to 3 dB compression. However, 3 dB compression can be assumed close to saturation, and the comparison results indicate that OLMBA is able to squeeze more power out, with better maximum and backed-off efficiency. PNA-X differential IQ mode was used in the additional BA measurements and that causes some phase fluctuation in the AM-PM results of the stand-alone BA. Figure 10 also contains simulated results of the OLMBA, which show roughly 1 dB more gain and slightly higher PAE. Overall, the simulations match well with the measurements.

Figure 9. (a) Diagram and (b) micrograph of the stand-alone quadrature balanced PA.

Figure 10. Simulated OLMBA results and comparison between OLMBA and stand-alone quadrature balanced PA power sweep measurement results, measured at 26 GHz. Light blue dashed lines indicate simulated OLMBA results and the diamond shapes indicate 1-dB compression points.

Table 2. CW measurements results comparison between stand-alone BA and OLMBA at 26 GHz

Modulated measurements and comparison to the state-of-the-art

Modulated measurements were conducted using the same bias settings that were used with continuous wave measurements presented in the previous section. Three waveforms were chosen as test signals: 100 and 400 MHz 64-QAM 3GPP/NR frequency range 2 (FR2) waveforms (10.9 dB peak to average power ratio with 1e-3 peak probability) and a 0.6 Gb/s (120 MHz) single carrier (SC) 64-QAM signal (0.35 roll-off factor). The signals were generated with Keysight arbitrary waveform generator (Keysight M8190A) and with Keysight programmable signal generator (Keysight E8257D). The output channel power and adjacent channel leakage ratio (ACLR) were measured with Keysight UXA (N9040B) signal analyzer. Error vector magnitude (EVM) was captured using vector signal analyzer (VSA) software. Same as with previous measurements, the sweeps were controlled and data were captured with MATLAB. Measurement setup EVM and ACLR before the PA under test were measured to be 2.3% (−32.8 dB) and −42 dBc, respectively. Block diagram of the modulated measurement setup is shown in Fig. 6(b).

3GPP defined limits for ACLR and EVM were used as threshold values for reporting the average output power with modulated signals. The limits are −28c dB and 8% (−21.9 dB) for ACLR and EVM, respectively [5]. ACLR measurements with three different signals are presented in Fig. 11(a) and EVM sweeps in Fig. 11(b). ACLR can be seen to be more limiting, especially with 400 MHz signal. Measured at 26 GHz, the PA reaches 11.4 dBm P _avg with 100 MHz 64-QAM OFDM signal and 4.9 dBm with 400 MHz 64-QAM OFDM signal. With SC signal, OLMBA achieves 14 dBm P _avg with good margin of linearity. Results with 100 MHz OFDM signal are also summarized in Table 1, where the PA shows steady performance throughout the tested frequencies, with average power fulfilling the linearity specifications staying within 1 dB. Figure 11 contains also stand-alone BA measurement results with 100 MHz 64-QAM OFDM signal for reference. Stand-alone BA can be seen to have better linearity in deep back-off but reaches the same output power with the linearity threshold. The differences between BA and OLMBA modulated performance are further examined in Table 3. The comparison is done at two average output power levels set 3 dB apart. OLMBA provides better efficiency in backed-off power levels.

Figure 11. (a) ACLR and (b) EVM measurement results at 26 GHz.

Table 3. Modulated signal performance comparison between stand-alone BA and OLMBA at 26 GHz with 100 MHz 64-QAM OFDM signal

Measurement results at 26 GHz are compared to other state-of-the-art advanced PA solutions in Table 4. With CW results, the proposed PA delivers comparable output power, gain, and output compression point. With modulated signals, the proposed PA provides comparable results, especially considering that most of the others were not tested using 5G NR signals. It should be noted that SC 64-QAM signal can have 4–5 dB lower PAPR compared to the OFDM 64-QAM used in this work [Reference Wang, Asbeck and Fager12, Reference Huang, Mannem, Li, Jung, Huang and Wang32].

Table 4. Comparison to the state-of-the-art